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10.HRecommendation G.722:
07 kHz AUDIOCODING WITHIN 64 KBIT/S
1. General
1.1HScope and outline description
HThis Recommendation describes the characteristics of an audio (50 to
7 000 Hz) coding system which may be used for a variety of higher quality
speech applications. The coding system uses subband adaptive differential pulse code
modulation (SBADPCM) within a bit rate of 64 kbit/s. The system is henceforth
referred to as 64 kbit/s (7 kHz) audio coding. In the SBADPCM technique used,
the frequency band is split into two subbands (higher and lower) and the signals
in each subband are encoded using ADPCM. The system has three basic modes of
operation corresponding to the bit rates used for 7 kHz audio coding: 64, 56 and
48 kbit/s. The latter two modes allow an auxiliary data channel of 8 and 16
kbit/s respectively to be provided within the 64 kbit/s by making use of bits from
the lower subband.
HFigure 1/G.722 identifies the main functional parts of the 64 kbit/s (7
kHz) audio codec as follows:
Hi) 64 kbit/s (7 kHz) audio encoder comprising:
a transmit audio part which converts an audio signal to a
Huniform digital signal which is coded using 14 bits with
H16kHz sampling;
H
a SBADPCM encoder which reduces the bit rate to 64kbit/s.
Hii)
64 kbit/s (7 kHz) audio decoder comprising:
H
a SBADPCM decoder which performs the reverse operation to the
encoder noting that the effective audio coding bit rate at the
input of the decoder can be 64, 56 or 48 kbit/s depending on
the mode of operation;
H
a receive audio part which reconstructs the audio signal from
the uniform digital signal which is encoded using 14bits with
16 kHz sampling.
HThe following two parts, identified in Figure 1/G.722 for clarification,
will be needed for applications requiring an auxiliary data channel within the
64 kbit/s:
H
a data insertion device at the transmit end which makes use
of, when needed, 1 or 2 audio bits per octet depending on the
mode of operation and substitutes data bits to provide an
auxiliary data channel of 8 or 16 kbit/s respectively;
H
a data extraction device at the receive end which determines
the mode of operation according to a mode control strategy and
extracts the data bits as appropriate.
Hss 1.2 contains a functional description of the transmit and receive
audio parts, ss 1.3 describes the modes of operation and the implication of
inserting data bits on the algorithms, whilst ssss 1.4 and 1.5 provide the functional
descriptions of the SBADPCM encoding and decoding algorithms respectively. ss
1.6 deals with the timing requirements. ss 2 specifies the transmission
characteristics of the 64 kbit/s (7 kHz) audio codec and of the transmit and receive audio
parts, ssss 3 and 4 give the principles of the SB ADPCM encoder respectively
whilst ssss 5 and 6 specify the computational details of the Quadrature Mirror
Filters (QMF) and of the ADPCM encoders and decoders respectively.
HNetworking aspects and test sequences are addressed in Appendices I and
II respectively to this Recommendation.
HRecommendation G.725: "Systems aspects for the use of the 7 kHz audio
codec within 64 kbit/s" contains specifications for inchannel handshaking
procedures for terminal identification and for mode control strategy, including
interworking with existing 64 kbit/s PCM terminals.
1.2HFunctional description of the audio parts
HFigure 2/G.722 shows a possible arrangement of audio parts in a
64kbit/s (7 kHz) audio coding terminal. The microphone, preamplifier, power amplifier
and loudspeaker are shown simply to identify the audio parts and are not
considered further in this Recommendation.
HIn order to facilitate the measurement of the transmission
characteristics as specified in ss 2, test points A and B need to be provided as shown. These
test points may either be for test purposes only or, where the audio parts are
located in different units from the microphone, loudspeaker etc..., correspond to
physical interfaces.
HThe transmit and receive audio parts comprise either the following
functional units or any equivalent items satisfying the specifications of ss 2:
Hi) transmit:
H
an input level adjustment device,
H
an input antialiasing filter,
H
a sampling device operating at 16 kHz,
H
an analoguetouniform digital converter with 14 bits and K#LL$MM$Nwith 16 kHz sampling;
Hii)
receive:
H
a uniform digitaltoanalogue converter with 14 bits and J(#KK#LL$Mwith 16 kHz sampling,
H
a reconstructing filter which includes x/sin x correction,
H
an output level adjustment device.
1.3HPossible modes of operation and implications of inserting data
HThe three basic possible modes of operation which correspond to the bit
rates available for audio coding at the input of the decoder are defined in
Table 1/G.722.
HITABLE 1/G.722
HO
H?Basic possible modes of operationă
Mode 7 kHz audio coding bit rate Auxiliary data channel bit rate
1 64 kbit/s 0 kbit/s
2 56 kbit/s 8 kbit/s
3 48 kbit/s 16 kbit/s
HSee Appendix 1 for examples of applications using one or several of
these modes and for their corresponding subjective quality.
HThe 64 kbit/s (7 kHz) audio encoder uses 64 kbit/s for audio coding at
all times irrespective of the mode of operation. The audio coding algorithm has
been chosen such that, without sending any indication to the encoder, the least
significant bit or two least significant bits of the lower subband may be used
downstream from the 64 kbit/s (7 kHz) audio encoder in order to substitute the
auxiliary data channel bits. However, to maximize the audio performance for a
given mode of operation, the 64 kbit/s (7 kHz) audio decoder must be optimized to
the bit rate available for audio coding. Thus, this Recommendation describes three
variants of the SBADPCM decoder and, for applications requiring an auxiliary
data channel, an indication must be forwarded to select in the decoder the variant
appropriate to the mode of operation. Figure 1/G.722 illustrates the
arrangement. It should be noted that the bit rate at the input of the 64 kbit/s (7 kHz)
audio decoder is always 64 kbit/s but comprising 64, 56 or 48 kbit/s for audio
coding depending on the mode of operation. From an algorithm viewpoint, the variant
used in the SBADPCM decoder can be changed in any octet during the
transmission. When no indication about the mode of operation is forwarded to the decoder,
the variant corresponding to Mode 1 should be used.
HA mode mismatch situation, where the variant used in the 64 kbit/s
(7kHz) audio decoder for a given octet does not correspond to the mode of operation,
will not cause misoperation of the decoder. However, to maximize the audio
performance, it is recommended that the mode control strategy adopted in the data
extraction device should be such as to minimize the duration of the mode mismatch.
Appendix I gives further information on the effects of a mode mismatch. To
ensure compatibility between various types of 64 kbit/s (7 kHz) audio coding
terminals, it is recommended that, as a minimum, the variant corresponding to Mode 1
operation is always implemented in the decoder.
HThe mode control strategy could be derived from the auxiliary data
channel protocol (see draft Recommendation G.725).
1.4HFunctional description of the SBADPCM encoder
HFigure 3/G.722 is a block diagram of the SBADPCM encoder. A functional
description of each block is given below in ssss 1.4.1 to 1.4.4.
1.4.1HTransmit quadrature mirror filters (QMFs)
HThe transmit QMFs comprise two linearphase nonrecursive digital
filters which split the frequency band 0 to 8 000 Hz into two subbands: the lower subband (0 to 4 000 Hz) and the higher subband (4 000 to 8 000 Hz). The input to
the transmit QMFs, xin, is the output from the transmit audio part and is
sampled at 16 kHz. The outputs, xL and xH, for the lower and higher subbands
respectively, are sampled at 8 kHz.
1.4.2HLower subband ADPCM encoder
HFigure 4/G.722 is a block diagram of the lower subband ADPCM encoder.
The lower subband input signal, xL after subtraction of an estimate, sL, of the
input signal produces the difference signal, eL. An adaptive 60level nonlinear
quantizer is used to assign six binary digits to the value of the difference
,signal to produce a 48 kbit/s signal, IL.
HIn the feedback loop, the two least significant bits of IL are deleted
to produce a 4bit signal ILt, which is used for the quantizer adaptation and
applied to a 15level inverse adaptive quantizer to produce a quantized difference
signal, dLt. The signal estimate, sL is added to this quantized difference
signal to produce a reconstructed version, rLt, of the lower subband input signal.
Both the reconstructed signal and the quantized difference signal are operated
upon by an adaptive predictor which produce the estimate, sL, of the input signal,
thereby completing the feedback loop.
H4bit operation, instead of 6bit operation, in the feedback loops of
both the lower subband ADPCM encoder, and the lower subband ADPCM decoder allows
the possible insertion of data in the two least significant bits as described in
ss 1.3 without causing misoperation in the decoder. Use of a 60 level quantizer
(instead of 64level) ensures the pulse density requirements as described in
Recommendation G.802 are met under all conditions and in all modes of operation.
1.4.3HHigher subband ADPCM encoder
HFigure 5/G.722 is a block diagram of the higher subband ADPCM encoder.
The higher subband input signal, xH after subtraction of an estimate, sH, of
the input signal, produces the difference signal, eH. An adaptive 4level non
linear quantizer is used to assign two binary digits to the value of the difference
signal to produce a 16 kbit/s signal, IH.
HAn inverse adaptive quantizer produces a quantized difference signal,
dH, from these same two binary digits. The signal estimate, sH, is added to this
quantized difference signal to produce a reconstructed version, rH, of the higher
subband input signal. Both the reconstructed signal and the quantized
difference signal are operated upon by an adaptive predictor which produces the estimate,
sH, of the input signal, thereby completing the feedback loop.
1.4.4HMultiplexer
HThe multiplexer (MUX) shown in Figure 3/G.722 is used to combine the
signals, IL and IH, from the lower and higher subband ADPCM encoders respectively
into a composite 64 kbit/s signal, I, with an octet format for transmission.
HThe output octet format, after multiplexing, is as follows:
HIH1 IH2 IL1 IL2 IL3 IL4 IL5 IL6
where IH1 is the first bit transmitted, and where IH1 and IL1 are the most
significant bits of IH and IL respectively, whilst IH2 and IL6 are the least
significant bits of IH and IL respectively.
1.5HFunctional description of the SBADPCM decoder
HFigure 6/G.722 is a block diagram of the SBADPCM decoder. A functional
description of each block is given below in ssss 1.5.1 to 1.5.4.
1.5.1HDemultiplexer
HThe demultiplexer (DMUX) decomposes the received 64 kbit/s octet
formatted signal, Ir, into two signals, ILr and IH, which form the codeword inputs to
the lower and higher subband ADPCM decoders respectively.
1.5.2HLower subband ADPCM decoder
HFigure 7/G.722 is a block diagram of the lower subband ADPCM decoder.
This decoder can operate in any of three possible variants depending on the
received indication of the mode of operation.
HThe path which produces the estimate, sL, of the input signal including
the quantizer adaptation, is identical to the feedback portion of the lower sub
band ADPCM encoder described in ss 1.4.2. The reconstructed signal, rL, is
produced by adding to the signal estimate one of three possible quantized difference
signals, dL,6, dL,5 or dL,4 (= dLt see note), selected according to the
received indication of the mode of operation. For each indication, Table2/G.722 shows
the quantized difference signal selected, the inverse adaptive quantizer used
and the number of least significant bits deleted from the input codeword.
HITABLE 2/G.722
HO
H=Lower subband ADPCM decoder variantsă
Received Quantized Inverse Number of least
indication difference adaptive significant bits
of mode of signal quantizer deleted from input
operation selected used codeword, ILr
Mode 1 dL,6 60level 0
Mode 2 dL,5 30level 1
Mode 3 dL,4 15level 2
Note For clarification purposes, all three inverse quantizers have been
indicated in the upper portion of Figure 7/G.722. In an optimized implementation, the
signal dLt, produced in the predictor loop, could be substituted for dL,4.
1.5.3HHigher subband ADPCM decoder
HFigure 8/G.722 is a block diagram of the higher subband ADPCM decoder.
This decoder is identical to the feedback portion of the higher subband ADPCM
encoder described in ss 1.4.3, the output being the reconstructed signal, rH.
1.5.4HReceive QMFs
HThe receive QMFs shown in Figure 6/G.722 are two linearphase non
recursive digital filters which interpolate the outputs, rL and rH, of the lower and
higher subband ADPCM decoders from 8 kHz to 16 kHz and which then produce an
output, xout, sampled at 16 kHz which forms the input to the receive audio parts.
HExcluding the ADPCM coding processes, the combination of the transmit
and the receive QMFs has an impulse response which closely approximates a simple
delay whilst, at the same time, the aliasing effects associated with the 8 kHz
subsampling are cancelled.
1.6HTiming requirements
H64 kHz bit timing and 8 kHz octet timing should be provided by the
network to the audio decoder.
HFor a correct operation of the audio coding system, the precision of the
16 kHz sampling frequencies of the A/D and D/A converters must be better than +
50.10é6.
2.HTransmission characteristics
2.1HCharacteristics of the audio ports and the test points
,Ԍ
HFigure 2/G.722 indicates the audio input and output ports and the test
points (A and B). It is for the designer to determine the characteristics of the
audio ports and the test points (i.e. relative levels, impedances, whether
balanced or unbalanced). The microphone, preamplifier, power amplifier and
loudspeaker should be chosen with reference to the specifications of the audio parts: in
particular their nominal bandwidth, idle noise and distortion.
HIt is suggested that input and output impedances should be high and low,
respectively, for an unbalanced termination whilst for a balanced termination
these impedances should be 600 ohms. However, the audio parts should meet all
audio parts specifications for their respective input and output impedance
conditions.
2.2HOverload point
HThe overload point for the analoguetodigital and digitaltoanalogue
converters should be + 9 dBmO + 0.3 dB. This assumes the same nominal speech
level (see Recommendation G.232) as for 64 kbit/s PCM but with a wider margin for
the maximum signal level which is likely to be necessary with conference
arrangements. The measurement method of the overload point is under study.
2.3HNominal reference frequency
HWhere a nominal reference frequency of 1 000 Hz is indicated below, the
actual frequency should be chosen equal to 1 020 Hz. The frequency tolerance
should be +2 to 7 Hz.
2.4HTransmission characteristics of the 64 kbit/s (7 kHz) audio codec
HThe values and limits specified below should be met with a 64 kbit/s
(7kHz) audio encoder and decoder connected backtoback. For practical reasons,
the measurements may be performed in a looped configuration as shown in
Figure9a/G.722. However, such a looped configuration is only intended to simulate an
actual situation where the encoder and decoder are located at the two ends of a
connection.
HThese limits apply to operation in Mode 1.
2.4.1HNominal bandwidth
HThe nominal 3 dB bandwidth is 50 to 7 000 Hz.
2.4.2HAttenuation/frequency distortion
HThe variation with frequency of the attenuation should satisfy the
limits shown in the mask of Figure 10/G.722. The nominal reference frequency is 1 000
Hz and the test level is 10 dBmO.
2.4.3HAbsolute group delay
HThe absolute group delay, defined as the minimum group delay for a
sinewave signal between 50 and 7 000 Hz, should not exceed 4 ms. The test level is
10 dBmO.
2.4.4HIdle noise
HThe unweighted noise power measured in the frequency range 50 to
7000Hz with no signal at the input port (test point A) should not exceed 66dBmO.
When measured in the frequency range 50 Hz to 20 kHz the unweighted noise power
should not exceed 60 dBmO.
2.4.5HSingle frequency noise
HThe level of any single frequency (in particular 8 000 Hz, the sampling
frequency and its multiples), measured selectively with no signal at the input
port (test point A) should not exceed 70 dBmO.
2.4.6HSignaltototal distortion ratio
HUnder study.
2.5HTransmission characteristics of the audio parts
HWhen the measurements indicated below for the audio parts are from audiotoaudio, a looped configuration as shown in Figure 9b/G.722 should be used.
The audio parts should also meet the specifications of ss 2.4 with the measurement
configuration of Figure 9b/G.722.
2.5.1HAttenuation/frequency response of the input antialiasing filter
HThe inband and outofband attenuation/frequency response of the input
antialiasing filter should satisfy the limits of the mask shown in
Figure11/G.722. The nominal reference frequency is 1 000 Hz and the test level for the inband characteristic is 10 dBmO. Appropriate measurements should be made to
check the outofband characteristic taking into account the aliasing due to the 16
kHz sampling.
2.5.2HAttenuation/frequency response of the output reconstructing filter
HThe inband and outofband attenuation/frequency response of the output
reconstructing filter should satisfy the limits of the mask shown in Figure
12/G.722. The nominal reference frequency is 1 000 Hz and the test level for the inband characteristic is 10 dBmO. Appropriate measurements should be made to
check the outofband characteristic taking into account the aliasing due to the 16
kHz sampling. The mask of Figure 12/G.722 is valid for the whole of the receive
audio part including any pulse amplitude modulation distortion and x/sin x
correction.
2.5.3HGroup delay distortion with frequency
HThe group delay distortion, taking the minimum value of group delay as a
reference, should satisfy the limits of the mask shown in Figure 13/G.722.
2.5.4HIdle noise for the receive audio part
HThe unweighted noise power of the receive audio part measured in the
frequency range 50 to 7 000 Hz with a 14bit allzero signal at its input should
not exceed 75 dBmO.
2.5.5HSignaltototal distortion ratio as a function of input level
HWith a sine wave signal at a frequency excluding simple harmonic
relationships with the 16 kHz sampling frequency, applied to test point A, the ratio of
signaltototal distortion power as a function of input level measured
unweighted in the frequency range 50 to 7 000 Hz at test point B, should satisfy the
limits of the mask shown in Figure 14/G.722. Two measurements should be performed,
one at a frequency of about 1 kHz and the other at a frequency of about 6 kHz.
2.5.6HSignaltototal distortion ratio as a function of frequency
HWith a sine wave signal at a level of 10 dBmO applied to test point A,
,the ratio of signaltototal distortion power as a function of frequency
measured unweighted in the frequency range 50 to 7 000 Hz at test point B should
satisfy the limits of the mask shown in Figure 15/G.722.
2.5.7HVariation of gain with input level
HWith a sine wave signal at the nominal reference frequency of 1 000 Hz,
but excluding the submultiple of the 16 kHz sampling frequency, applied to test
point A, the gain variation as a function of input level relative to the gain at
an input level of 10 dBmO measured selectively at test point B, should satisfy
the limits of the mask shown in Figure 16/G.722.
2.5.8HIntermodulation
HUnder study.
2.5.9HGo/return crosstalk
HThe crosstalk from the transmit direction to the receive direction
should be such that, with a sine wave signal at any frequency in the range 50 to 7
000 Hz and at a level of +6 dBmO applied to test point A, the crosstalk level
measured selectively at test point B should not exceed 64 dBmO. The measurement
should be made with a 14bit allzero digital signal at the input to the receive
audio part.
HThe crosstalk from the receive direction to the transmit direction
should be such that, with a digitally simulated sine wave signal at any frequency in
the range of 50 to 7 000 Hz and a level of +6 dBmO applied to the input of the
receive audio part, the crosstalk level measured selectively and with the
measurement made digitally at the output of the transmit audio part should not exceed
64 dBmO. The measurement should be made with no signal at test point A, but
with the test point correctly terminated.
2.6HTranscoding to and from 64 kbit/s PCM
HFor compatibility reasons with 64 kbit/s PCM, transcoding between
64kbit/s (7 kHz) audio coding and 64 kbit/s PCM should take account of the relevant
specifications of Recommendations G.712, G.713 and G.714. When the audio signal
is to be heard through a loudspeaker, more stringent specifications may be
necessary. Further information may be found in Appendix 1.
3.HSBADPCM encoder principles
HA block diagram of the SBADPCM encoder is given in Figure 3/G.722.
Block diagrams of the lower and higher subband ADPCM encoders are given
respectively in Figures 4/G.722 and 5/G.722.
HMain variables used for the descriptions in ssss 3 and 4 are summarized
in Table 3/G.722. In these descriptions, index (j) indicates a value
corresponding to the current 16 kHz sampling interval, index (jl) indicates a value
corresponding to the previous 16 kHz sampling interval, index (n) indicates a value
corresponding to the current 8 kHz sampling interval, and index (nl) indicates a
value corresponding to the previous 8 kHz sampling interval. Indices are not
used for internal variables, i.e. those employed only within individual
computational blocks.
3.1HTransmit QMF
HA 24coefficient QMF is used to compute the lower and higher subband
signal components. The QMF coefficient values, hi, are given in Table 4/G.722.
HThe output variables, xL(n) and xH(n), are computed in the following
way:
HxL(n) = xA + xB (1)
xH(n) = xA xB (2)
where
11
HxA = $ h2i.xin(j 2i) (3)
i=0
11
HxB = $ h2i+1.xin(j 2i 1) (4)
i=0
3.2HDifference signal computation
HThe difference signal, eL(n) and eH(n), are computed by subtracting
predicted values, sL(n) and sH(n), from the lower and higher subband input values,
xL(n) and xH(n):
HeL(n) = xL(n) sL(n) (5)
HeH(n) = xH(n) sH(n) (6)
HITABLE 3/G.722
HO
H0Variables used in the SBADPCM encoder and decoder descriptionsă
Note Variables used exclusively within one section are not listed. Subscripts
L and H refer to lower subband and higher subband values. Subscript Lt denotes
values generated from the truncated 4bit codeword as opposed to the
nontruncated 6bit (encoder) or 6/5/ or 4bit (decoder) codewords.
HITABLE 4/G.722
HO
H:Transmit and receive QMF coefficient valuesă
3.3HAdaptive quantizer
HThe difference signals, eL(n) and eH(n), are quantized to 6 and 2 bits
for the lower and higher subbands respectively. Tables 5/G.722 and 6/G.722 give
the decision levels and the output codes for the 6 and 2bit quantizers
respectively. In these tables, only the positive decision levels are indicated, the
negative levels can be determined by symmetry. mL and mH are indices for the
quantizer intervals. The interval boundaries, LL6, LU6, HL and HU, are scaled by
computed scale factors, L(n) and H(n) (see ss 3.5). Indices, mL and mH, are then
determined to satisfy the following:
HLL6(mL).L(n) eL(n) < LU6(mL).L(n) (7)
HHL(mH).H(n) eH(n) < HU(mH).H(n) (8)
for the lower and higher subbands respectively.
HThe output codes, ILN and IHN, represent negative intervals, whilst the
output codes, ILP and IHP, represent positive intervals. The output codes, IL(n)
and IH(n), are then given by:
ILP(mL) , if eL (n) 0
HIL (n) = (9)
ILN(mL) , if eL (n) < 0
H
IHP(mH) , if eH (n) 0
IH (n) = (10)
IHN(mH) , if eH (n) < 0
for the lower and higher subbands respectively.
3.4HInverse adaptive quantizers
3.4.1HInverse adaptive quantizer in the lower subband ADPCM encoder
HThe lower subband output code, IL(n), is truncated by two bits to
produce ILt(n). The 4bit codeword, ILt(n), is converted to the truncated quantized
difference signal, dLt(n), using the QL4é1 output values of Table7/G.722, and
scaled by the scale factor, L(n):
HdLt(n) = QL4é1(ILt(n)).L(n).sgn(ILt(n)) (11)
where sgn (ILt(n)) is derived from the sign of eL(n) defined in equation 9.
HThere is a unique mapping, shown in Table 7/G.722, between four adjacent
6bit quantizer intervals and the QL4é1 output values. QL4é1(ILt(n)) is
determined in two steps: first determination of the quantizer interval index, mL,
corresponding to IL(n) from Table 5/G.722, and then determination of
QL4é1(mL) by reference to Table 7/G.722.
3.4.2HInverse adaptive quantizer in the higher subband ADPCM encoder
HThe higher subband output code, IH(n) is converted to the quantized
difference signal, dH(n), using the Q2é1 output values of Table 8/G.722 and
scaled by the scale factor, H(n):
HdH(n) = Q2é1(IH(n)).H(n).sgn(IH(n)) (12)
HITABLE 5/G.722
HO
H,Decision levels and output codes for the 6bit lower subband quantizeră
Note If a transmitted codeword for the lower subband signal has been
transformed, due to transmission errors to one of the four suppressed codewords
"0000XX", the received code word is set at "111111".
HO
HITABLE 6/G.722
HO
H+Decision levels and output codes for the 2bit higher subband quantizeră
mH HL HH IHN IHP
1 0 1.10156 01 11
2 1.10156 00 10
HITABLE 7/G.722
HO
HOutput values and multipliers for 6, 5 and 4bit lower subband
inverse quantizers
HITABLE 8/G.722
HO
H-Output values and multipliers for the 2bit higher subband quantizeră
mH Q2é1 WH
1 0.39453 0.10449
2 1.80859 0.38965
where sgn (IH(n)) is derived from the sign of eH(n) defined in equaltion (10),
and where Q2é1(IH(n)) is determined in two steps: first determine the quantizer
interval index, mH, corresponding to IH(n) from Table 6/G.722 and then determine
Q2é1(mH) by reference to Table 8/G.722.
3.5HQuantizer adaptation
HThis block defines L(n) and H(n), the scaling factors for the lower
and higher subband quantizers. The scaling factors are updated in the log domain
and subsequently converted to a linear representation. For the lower sub band,
the input is ILt(n), the codeword truncated to preserve the four most
significant bits. For the higher subband, the 2bit quantizer output, IH(n), is used
directly.
HFirstly the log scaling factors, L(n) and H(n), are updated as
follows:
L(n) = B. L(n1)WL(ILt(n1)) (13)
H(n) = B. H(n1)WH(IH(n1)) (14)
where WL(.) and Wy(.) are logarithmic scaling factor multipliers given in
Tables7/G.722 and 8/G.722, and B is a leakage constant equal to 127/128.
HThen the log scaling factors are limited, according to:
0 L(n) 9 (15)
0 H(n) 11 (16)
HFinally, the linear scaling factors are computed from the log scaling
factors, using an approximation of the inverse log2 function:
L(n) = 2( L(n)2)min (17)
H(n) = 2 H(n)min (18)
where min is equal to half the quantizer step size of the 14 bit analogueto
digital converter.
3.6HAdaptive prediction
3.6.1HPredicted value computations
HThe adaptive predictors compute predicted signal values, sL(n) and
sH(n), for the lower and higher subbands respectively.
HEach adaptive predictor comprises two sections: a secondorder section
that models poles, and a sixthorder section that models zeroes in the input
signal.
HThe second order pole sections (coefficients aL,i and aH,i) use the
quantized reconstructed signals, rLt(n) and rH(n), for prediction. The sixth order
zero sections (coefficients bL,i and bH,i) use the quantized difference signals,
dLt(n) and dH(n). The zerobased predicted signals, sLz(n) and sHz(n), are also
employed to compute partially reconstructed signals as described in ssĠ3.6.2.
4.HSBADPCM decoder principles
HA block diagram of the SBADPCM decoder is given in Figure 6/G.722 and
block diagrams of the lower and higher subband ADPCM decoders are given
respectively in Figures 7/G.722 and 8/G.722.
HThe input to the lower subband ADPCM decoder, ILr, may differ from IL
even in the absence of transmission errors, in that one or two least significant
bits may have been replaced by data.
4.1HInverse adaptive quantizer
4.1.1HInverse adaptive quantizer selection for the lower subband ADPCM I"Jdecoder
HAccording to the received indication of the mode of operation the number
of least significant bits which should be truncated from the input codeword ILr,
and the choice of the inverse adaptive quantizer are determined, as shown in
Table 2/G.722.
HFor operation in mode 1, the 6bit codeword, ILr(n), is converted to the
quantized difference, dL(n), according to QL6é1 output values of Table7/G.722,
and scaled by the scale factor, L(n):
HdL(n) = QL6é1(ILr(n)).L(n).sgn (ILr(n)) (39)
where sgn (ILr(n)) is derived from the sign of IL(n) defined in equation (9).
HSimilarly, for operations in mode 2 or mode 3, the truncated codeword
(by one or two bits) is converted to the quantized difference signal, dL(n),
according to QL5é1 or QL4é1 output values of Table 7/G.722 respectively.
HThere are unique mappings, shown in Table 7/G.722, between two or four
adjacent 6bit quantizer intervals and the QL5é1 or QL4é1 output values
respectively.
HIn the computations above, the output values are determined in two
steps: first determination of the quantizer interval index, mL, corresponding to
ILr(n) from Table 5/G.722, and then determination of the output values corresponding
to mL by reference to Table 7/G.722.
HThe inverse adaptive quantizer, used for the computation of the
predicted value and for adaptation of the quantizer and predictor, is described in ss
3.4.1, but with IL(n) replaced by ILr(n).
4.1.2HInverse adaptive quantizer for the higher subband ADPCM decoder
HSee ss 3.4.2.
4.2HQuantizer adaptation
HSee ss 3.5.
4.3HAdaptive prediction
4.3.1HPredicted value computation
HSee ss 3.6.1.
4.3.2HReconstructed signal computation
HSee ss 3.6.2.
HThe output reconstructed signal for the lower subband ADPCM decoder,
rL(n), is computed from the quantized difference signal, dL(n), as follows:
HrL(n) = SL(n) + dL(n) (40)
4.3.3HPole section adaptation
HSee ss 3.6.3.
4.3.4HZero section adaptation
HSee ss 3.6.4.
4.4HReceive QMF
HA 24coefficient QMF is used to reconstruct the output signal, xout(j),
from the reconstructed lower and higher subband signals, rL(n) and rH(n). The
QMF coefficient values, hi, are the same as those used in the transmit QMF and
are given in Table 4/G.722.
HThe output signals, xout(j) and xout(j+1), are computed in the following
way:
(41)
(42)
(43)
(44)
H:Attenuation/frequency response of the inpută
HFantialiasing filteră
H9Attenuation/frequency response of the outpută
H6reconstructing filter (including x/sinx correction)ă
H<Signaltototal distortion ratio as a ă
HEfunction of frequencyă
7I(3291)
CCITT\APIX\DOC\142E8.TXS
77I(3291)
CCITT\APIX\DOC\142E8.TXS
7 X
5.HComputational details for QMF
5.1HInput and output signals
HTable 9/G.722 defines the input and output signals for the transmit and
receive QMF. All input and output signals have 16bit wordlengths, which are
limited to a range of 16384 to 16383 in 2's complement notation. Note that the
most significant magnitude bit of the A/D output and the D/A input appears at the
third bit location in XIN and XOUT, respectively.
5.2HDescription of variables and detailed specification of subblocks
HThis section contains a detailed expansion of the transmit and receive
QMF. The expansions are illustrated in Figures 17/G.722 and 18/G.722 with the
internal variables given in Table 10/G.722, and the QMF coefficients given in Table
11/G.722. The wordlengths of the internal variables, XA, XB and WD, must be
equal to or greater than 24 bits (see Note 1). The other internal variables have a
minimum of 16 bit wordlengths. A brief functional description and the full
specification is given for each subblock.
HThe notations used in the block descriptions are as follows:
H>>n
denotes an nbit arithmetic shift right operation (sign H8"II"Jextension),
H+denotes arithmetic addition with saturation control which J(#KK#Lforces the result to minimum or maximum representable value ; <<=in case of
underflow or overflow, respectively,
Hdenotes arithmetic subtraction with saturation control which M$NN%Oforces the result to minimum or maximum representable value <==>in case of
underflow or overflow, respectively.
H*denotes arithmetic multiplication which can be performed with N%OO%Peither truncation or rounding,
H<denotes the "less than" condition as x < y; x is less than y,
H>denotes the "greater than" condition, as x > y; x is greater M$NN%Othan y,
H= denotes the substitution of righthand variable for lefthand N%OO%Pone.
Note 1 Some freedom is offered for the implementation of the accumulation
process in the QMF: the wordlengths of the internal variables can be equal to or
greater than 24 bits, and the arithmetic multiplications can be performed with
either truncation or rounding. It allows a simplified implementation on various
types of processors. The counterpart is that it excludes the use of digital test
sequence for the test of the QMF.